The past few years has witnessed the ever-increasing availability of relatively cheap, low power wireless data communication services, networks and devices, promising near wire speed transmission and reliability. One technology in particular, described in the IEEE Standard 802.11a (1999) and Draft IEEE Standard 802.11g (2002) High Rate PHY Supplements to the ANSI/IEEE Standard 802.11, 1999 edition, collectively incorporated herein fully by reference, has recently been commercialized with the promise of 54 Mbps effective bandwidth, making it a strong competitor to traditional wired Ethernet and the more ubiquitous “802.11b” or “WiFi” 11 Mbps mobile wireless transmission standard.
IEEE 802.11a and 802.11g or “802.11a/g” compliant transmission systems achieve their high data transmission rates using Orthogonal Frequency Division Modulation or OFDM encoded symbols mapped up to 64 QAM multicarrier constellation. Before final power amplification and transmission, the multicarrier OFDM symbol encoded symbols are converted into the time domain using Inverse Fast Fourier Transform techniques resulting in a relatively high-speed time domain signal with a large peak-to-average ratio (PAR). OFDM is also used in fixed broadband wireless access systems such as proposed in IEEE Standard 802.16a: Air Interface for Fixed Broadband Wireless Access Systems Part A: Systems between 2 and 1 GHz, Draft working document, February 2002, (“802.16a”) which is incorporated herein fully by reference.
The receiver portion of a wireless communications systems compliant with the aforementioned 802.11a/802.11g and 802.16a standards generally includes an RF receiving unit to handle RF downconversion and filtering of received signals in one or more stages and a baseband processing unit to demodulate OFDM encoded symbols bearing the data of interest. FIG. 2 illustrates a known receiver baseband processing unit 200, a channel model 201 and a corresponding encoding transmitter baseband processor 260 to conceptually illustrate the relationship between a received OFDM symbol Yn in relation to its originally transmitted counterpart OFDM symbol Xn and the pre FEC version {tilde over (X)}n, taking into account the channel impulse response h(t) and intervening noise v(t) (Additive White Gaussian Noise or AWGN in particular), assuming that the channel impulse response h(t) is shorter than the guard interval (using e.g. cyclic prefix). To ease understanding, conventional RF upconversion transmission, and downconversion interposing the baseband processing units 200 and 260, are omitted from FIG. 2. More specifically, the channel model 201 illustrates the following relationship:Yk,n=gn Hk,n·Xk,n+gnvk,n  (0)where Xk,n is the transmitted symbol of the kth sub-carrier during the nth OFDM symbol, Hk,n is the channel response corresponding to the kth subcarrier during the nth OFDM symbol, and Vk,n is the AWGN in the kth sub-carrier during the nth OFDM symbol. Here, gn represents the gain compensation for the baseband signal conveying the symbol Yn provided by an automatic gain control unit (not shown) to maximize the dynamic range and performance of analog to digital conversion. Here, assume gn=1 since, in the single receive path case, gn is common to all subcarriers and will not therefore affect viterbi decoder performance.
The digital form of each Yn OFDM symbol presented in the frequency domain is recovered after conventional analog-digital conversion and fast fourier transformation by the ADC 205 and the FFT unit 208 respectively of the received time domain analog baseband signal. Thereafter, demodulation (phase rotation) and Frequency domain EQualization (FEQ) through a FEQ unit 220 is performed on the Fast Fourier Transform (FFT) output to compute X (Equation 1 below) before sending it to the viterbi decoder 230 to estimate the most likely transmitted symbol {circumflex over (X)}n at the transmitter. That is, the FEQ 220 computes:
                                              ⁢                                            X              _                                      k              ,              n                                =                                                    Y                                  k                  ,                  n                                                                              H                  _                                                  k                  ,                  n                                                      =                                                                                H                    _                                                        k                    ,                    n                                    *                                ·                                  Y                                      k                    ,                    n                                                                                                                                                            H                      _                                                              k                      ,                      n                                                                                        2                                                                        (        1        )            where: 1) Xk,n is the demodulated symbol of the kth sub-carrier during the nth OFDM symbol; 2) Yk,n is the received noisy symbol of the kth subcarrier during the nth OFDM symbol; and 3) Hk,n is the channel estimate corresponding to the kth subcarrier during the nth OFDM symbol.
In the receiver of an 802.11a/g OFDM system, for example, this operation has to be performed for each of the 52 subcarriers of the OFDM symbol Yn. In a straightforward implementation, the computation requires ˜2 complex multiplications and one real division per sub-carrier. This is computationally more intensive than implementing the FFT that requires 96 complex multiplications per 52 carriers. In particular, the implementation of a division operation can be at least 3˜5 times more complex than that of a multiplication operation. Thus, it is desirable to eliminate the division or scaling operations required in the FEQ unit 220.
As shown in FIG. 2, the output of the FEQ unit 220 is typically sent to a Viterbi decoder 230 to recover or estimate, based on historical analysis of the traffic in the channels, the most likely transmitted sequence of data symbols or bits. Hard symbol by symbol decisions (via slicer 225) on the output of the FEQ can also be fed back to the channel estimator 210 to improve the channel estimates Hn.
In a conventional FEQ implementation such as shown in FIG. 2, the received symbol for each carrier, Yk,n is divided by its corresponding channel estimate, Hk,n that can vary from sub-carrier to sub-carrier in a frequency selective fading environment. Thus, the expected signal to noise ratio per symbol may be varying from sub-carrier to sub-carrier and may deviate significantly from the average. This can reduce the performance of a conventional Viterbi decoder (such as decoder 230) that derives its optimal branch metrics assuming that all the symbols have the same expected Additive White Gaussian Noise (AWGN) noise power. Thus, it is desirable to incorporate the so called Channel State Information (CSI) derived from the channel estimates into the Viterbi decoder metrics to minimize this effect.
It would also be desirable to simplify the configuration of the viterbi decoder and branch metric computation in particular without increasing complexity over conventional designs while retaining these CSI performance benefits.
Any OFDM systems operating in fading channels, including 802.11a/g and 802.16a communications systems, are susceptible to wireless channel fading and multipath interference, particularly in mobile deployments where intervening terrain and buildings block or alter line-of-sight reception. To combat this, known receiver designs may employ plural geographically dispersed antennae in the hopes that at least one of the antennae can avoid the characteristic multipath and fading. One common type of plural antennae receiver is called a selection diversity receiver or simply a diversity receiver, and includes logic to ascertain which one of the RF receive pathways is receiving a “best” version of an incident signal (based on e.g. Received Signal Strength Indication (RSSI) analysis), and will select this “best” signal for downconversion and demodulation decoding while disregarding the remaining received signals. Typically, this involves switching the RF receive pathway or channel bearing this “best signal” into communication with the RF downconversion and baseband processing units while isolating the remaining pathways. Performance gains here remain dependent on individual antennae configuration and placement, but is invariably improved over conventional single channel designs.
Another form of the plural antennae receiver offering significant SNR benefit is the Maximum Ratio Combining (MRC) receiver, in which all the plural received signals are utilized. In particular, the received symbols within the plural signals borne by the RF receive pathways are co-phased to provide coherent voltage addition and are individually weighted to provide optimal SNR. See e.g. T. S. Rappaport, Wireless Communications, Prentice Hall, Upper Saddle River, N.J. 1996, pages 326- 332 which is incorporated herein fully by reference. As such, the baseband processing unit, especially the FEQ and Viterbi decoder of an MRC receiver is viewed as being especially complex, and so any reduction in such complexity, especially where performance is not affected or even is improved, is desirable.